This invention is in the field of digital communications, and is more specifically directed to power reduction techniques in communications involving multiple frequency bands.
An important and now popular modulation standard for digital subscriber line (DSL) communications is Discrete Multitone (DMT). According to DMT technology, the available spectrum is subdivided into many subchannels (e.g., 256 subchannels of 4.3125 kHz). Each subchannel is centered about a carrier frequency that is phase and amplitude modulated, typically by Quadrature Amplitude Modulation (QAM), in which each symbol value is represented by a point in the complex plane; the number of available symbol values depends, of course, on the number of bits in each symbol. During initialization of a DMT communications session, the number of bits per symbol for each subchannel (i.e., the “bit loading”) is determined according to the noise currently present in the transmission channel at each subchannel frequency and according to the transmit signal attenuation at that frequency. For example, relatively noise-free subchannels may communicate data in ten-bit to fifteen-bit symbols corresponding to a relatively dense QAM constellation (with short distances between points in the constellation), while noisy channels may be limited to only two or three bits per symbol (to allow a greater distance between adjacent points in the QAM constellation). Indeed, some subchannels may not be loaded with any bits, because of the noise and attenuation in those channels. In this way, DMT maximizes the data rate for each subchannel for a given noise condition, permitting high speed access to be carried out even over relatively noisy twisted-pair lines.
FIG. 1 illustrates the data flow in conventional DSL communications, for a given direction (e.g., downstream, from a central office “CO” to customer premises equipment “CPE”). Typically, each DSL transceiver (i.e., both at the CO and also in the CPE) includes both a transmitter and a receiver, so that data is also communicated in the opposite direction over transmission channel LP according to a similar DMT process. As shown in FIG. 1, the input bitstream that is to be transmitted, typically a serial stream of binary digits in the format as produced by the data source, is applied to constellation encoder 11 in a transmitting modem 10. Constellation encoder 11 fundamentally groups the bits in the input bitstream into multiple-bit symbols that are used to modulate the DMT subchannels, with the number of bits in each symbol determined according to the bit loading assigned to its corresponding subchannel, based on the characteristics of the transmission channel as mentioned above. Encoder 11 may also include other encoding functions, such as Reed-Solomon or other forward error correction coding, trellis coding, turbo or LDPC coding, and the like. The symbols generated by constellation encoder 11 correspond to points in the appropriate modulation constellation (e.g., QAM), with each symbol associated with one of the DMT subchannels. Following constellation encoder 11, shaping function 12 derives a clip prevention signal included in the encoded signals to be modulated, to reduce the peak-to-average ratio (PAR) as transmitted as described in copending application Ser. No. 10/034,951, filed Dec. 27, 2001, published on Nov. 28, 2002 as U.S. Patent Application Publication No. 2002/0176509, incorporated herein by this reference.
The encoded symbols are applied to inverse Discrete Fourier Transform (IDFT) function 13, which associates each symbol with one subchannel in the transmission frequency band, and generates a corresponding number of time domain symbol samples according to the Fourier transform. As known in the art, cyclic insertion function 14 appends a cyclic prefix or suffix, or both, to the modulated time-domain samples from IDFT function 13, and presents the extended block of serial samples to parallel-to-serial converter 15. In ADSL2+ and VDSL, cyclic prefix and suffix insertion, and transmitter windowing, are combined into a single cyclic insertion function 14, which preferably operates on the modulated data in parallel form as shown; in ADSL, cyclic insertion function 14 preferably follows serial-to-parallel conversion, and simply prepends a selected number of sample values from the end of the block to the beginning of the block. Following conversion of the time-domain signal into a serial sequence by converter 15, and such upsampling (not shown) as appropriate, digital filter function 16 then processes the digital datastream in the conventional manner to remove image components and voice band or ISDN interference. The filtered digital datastream signal is then converted into the analog domain by digital-to-analog converter 17. After conventional analog filtering and amplification (not shown), the resulting DMT signal is transmitted over a channel LP, over some length of conventional twisted-pair wires, to a receiving DSL modem 20, which, in general, reverses the processes performed by the transmitting modem to recover the input bitstream as the transmitted communication.
At receiving DSL modem 20, analog-to-digital conversion 22 then converts the filtered analog signal into the digital domain, following which conventional digital filtering function 23 is applied to augment the function of pre-conversion receiver analog filters (not shown). A time domain equalizer (TEQ) (not shown) may apply a finite impulse response (FIR) digital filter that effectively shortens the length of the impulse response of the transmission channel LP. Serial-to-parallel converter 24 converts the datastream into a number of samples (2N) for application to Discrete Fourier Transform (DFT) function 27, after removal of the cyclic extension from each received block in function 25. At DFT function 27, the modulating symbols at each of the subchannel frequencies are recovered by reversing the IDFT performed by function 12 in transmission. The output of DFT function 27 is a frequency domain representation of the transmitted symbols multiplied by the frequency-domain response of the effective transmission channel. Frequency-domain equalization (FEQ) function 28 divides out the frequency-domain response of the effective channel, recovering the modulating symbols. Constellation decoder function 29 then resequences the symbols into a serial bitstream, decoding any encoding that was applied in the transmission of the signal and producing an output bitstream that corresponds to the input bitstream upon which the transmission was based. This output bitstream is then forwarded to the client workstation, or to the central office network, as appropriate for the location.
The DMT communications process, such as shown in FIG. 1, provides excellent transmission data rates over modest communications facilities such as twisted-pair wires. However, multicarrier methods such as IDFT modulation can result in a high peak-to-average ratio (PAR) of the signal amplitudes. The PAR is defined as the ratio of the peak sample power level to the average power level over a sequence of samples. Relatively large peaks in the modulated time-domain signal can occur because the DMT modulated signal is the sum of many independent component signals. In modern DMT modulation, for example as used in DSL communications, the number of summed subchannel signals is sufficient that the well-known central limit theorem applies, and that therefore the amplitude of the IDFT-modulated time domain signal has a Gaussian-like probability distribution function.
FIG. 2 illustrates an example of a typical Gaussian probability distribution function as resulting from modern multichannel DSL communications, in which the amplitudes are distributed around a central mean with a deviation σ. While the ideal transmitter must be capable of driving all possible amplitudes in this distribution so that no information is lost, some clipping of peak amplitudes is necessitated in practical transmitters. Conventional communications standards are quite stringent, however, specifying amplitude clipping at the transmitter is not to exceed 10−7, to minimize loss of information. This requirement requires that the maximum amplitude available in the Gaussian probability distribution is about five times the deviation σ. In practice, therefore, the conventional DMT transmitter must have a line driver power supply voltage (“rail”) at voltage Vs (FIG. 1), at 5σ from the mean. The dynamic range required of conventional DMT transmitters to provide these high amplitudes while maintaining close resolution, particularly in the digital-to-analog conversion function, is very stringent.
As mentioned in Henkel, et al., “Another Application for Trellis Shaping: PAR Reduction for DMT (OFDM)”, Transactions on Communications, Vol. 48, No. 5 (IEEE, Sep. 2000), pp. 1471-76, similar concerns regarding PAR arise in orthogonal frequency-division multiplexing (OFDM) communications technologies, which are beginning to be used in applications other than DSL communications, including wireless telephony and wireless networks.
The high PAR for conventional DMT signals presents significant constraints on the transmission circuitry, and can greatly complicate the analog circuitry required for high fidelity transmission. For example, a high PAR requires a large dynamic range at the inputs of digital-to-analog and analog-to-digital converters, necessitating a large number of bits of resolution, thus greatly increasing the cost and complexity of these functions. Filters and amplifiers must also become more complex and costly in order to handle both the high peak amplitudes and also the resolution required for the vast majority of the samples having lower amplitude. In addition, the high PAR results in much higher power consumption in the communications circuits, further increasing the cost of the circuits and systems used for DMT transmission and receipt, particularly those circuits often referred to as the analog front end (AFE). Various techniques for reducing the PAR of DMT signals are known in the art, particularly in the DSL context. Examples of these techniques are described in copending application Ser. No. 10/034,951, filed Dec. 27, 2001, published on Nov. 28, 2002 as U.S. Patent Application Publication No. 2002/0176509; and in Gatherer and Polley, “Controlling clipping probability in DMT transmission”, Proceedings of the Asilomar Conference on Signals, Systems, and Computers, (1997), pp. 578-584, both incorporated herein by this reference.
After a DSL communications link has been established, for example between a central office (CO) and customer premises equipment (CPE), substantial periods of time may elapse in which no information traffic is being carried, but for which the active link is still to be maintained. These idle periods are referred to in the art as “quiescent” periods. Indeed, given the “always on” nature of modern broadband Internet access services, quiescent periods typically dominate active communication time for many users.
To reduce power consumption in DSL transceiver circuitry, particularly on the transmit side, it would be desirable to have a quiescent operating mode in which the average transmit power (e.g., the mean amplitude of, and all other points in, the Gaussian distribution of FIG. 1) is reduced for links that are in their idle, or quiescent, state. However, as known in the art, the initialization and “training” sequence for a new DSL communications link is based upon the signal-to-noise ratios, over frequency, of that new link, as affected by crosstalk interference from other communications links. If a given link is in its quiescent mode during the training of a new link in a neighboring facility, the training of the new link will set its transmit parameters (“bits” and “gains” for each DMT subchannel) based on the assumption that the low crosstalk from the first (idle) link will remain very low. However, once the first link exits its low power quiescent mode and again begins transmitting at full power, the interference on the newly-trained link will not have been comprehended in its training. The bit error rate of the newly-trained link will suffer, as a result.
Accordingly, it has been desirable to reduce the power supply voltage (“rail” voltage Vs of FIG. 1) of DSL transmitters in quiescent mode, thus reducing power dissipation at the transmitter, but without reducing the average power so that the training and initialization of neighboring links will be accurate. A quiescent mode that carries active signals at average power levels, but no information, would be attractive because this signal could be severely clipped to reduce the transmitter power supply rail voltage without loss of information. However, the complete absence of information in quiescent mode is too simplistic an assumption, because some higher layer protocols actually require communication of information during quiescent mode. Frequent entry into and exit from quiescent mode would therefore be necessary in this simple approach.
Various proposals for reducing the PAR in quiescent mode (referred to as “Q” mode, or “L2 mode”) were considered in adoption of the ADSL2 standard. Some of these proposals are summarized in Redfern, “A comparison of transparent physical layer Q-mode proposals”, submitted to Study Group 15 Question 4, ITU Telecommunications Standardization Sector, Document RN-083 (May 2001), incorporated herein by this reference. As described in that submission, however, these approaches require relatively complex operations at the transmitter, including XOR mapping and sign inversion; two of these approaches also involved phase rotation and post encoding operations that also required changes to the decoder. Because of concerns about the complexity and feasibility of these approaches, none were adopted for the standard.
Rather, conventional standard ADSL2 quiescent mode is achieved by reducing the average power from that used to transmit information-bearing DMT signals. This is accomplished by effectively re-initializing the DSL link, in which the transceivers negotiate a lower data rate in quiescent mode for downstream (CO to CPE) communications. This permits the CO transceiver to use a lower average fine gain level, thus reducing the average transmit power in this mode, and permitting implementation of a lower-power design for the CO transceiver.
However, this standard ADSL Q-mode approach has several drawbacks, as observed in practice. To effect this approach, the CO and CPE must each maintain two “bits and gains” tables for downstream communications—one for normal mode and another for Q mode; if Q mode is to be implemented also for upstream communications, then four such tables are required for each of the CO and CPE. This greatly increases the memory requirements for each transceiver, especially for ADSL2 and higher data rate DSL schemes in which as many as 4096 subchannels are used. In addition, entry into this Q mode requires the CO and CPE to calculate this second set of bits and gains for the large number of subchannels involved, and to exchange these values over the existing overhead channel, both of which consume computational and memory capacity. And since there is a separate bits and gains table for Q mode, and considering the possibility of variations in noise and environment in this mode, either the Q mode itself must support “bit swap” operations, or the transceivers must exit Q mode to effect this bit swap and then re-enter Q mode with the new bits and gains (causing unexpected interference with neighboring links). If the set of subchannels in Q mode is smaller than that in normal transmission, other maintenance operations such as updating frequency domain equalizers (FEQ) and the like cannot be done during Q mode. Framing parameters may also need to be reconfigured for this Q mode approach.